Periodic signal synchronization apparatus, systems, and methods

ABSTRACT

Apparatus, systems, and methods are disclosed that operate to generate a periodic output signal from a periodic input signal, obtain a plurality of samples of a phase difference between the output signal and the input signal, and to adjust a phase of the output signal based on the samples of the phase difference. Additional apparatus, systems, and methods are disclosed.

RELATED APPLICATION

This application is a Divisional of U.S. application Ser. No.11/788,442, filed Apr. 20, 2007, which is incorporated herein byreference in its entirety.

FIELD

This disclosure relates to periodic signal synchronization in electronicdevices.

BACKGROUND

Delay lock loop (DLL) circuits and phase lock loop (PLL) circuits areused to generate a periodic signal such as a clock signal based on aperiodic reference signal from, for example, an oscillator. Thegenerated clock signal should maintain a specific phase relationshipwith the reference signal to be synchronized. A DLL circuit or a PLLcircuit will adjust the phase of the generated clock signal to maintainthe desired phase relationship. DLL and PLL circuits are used, forexample, in high-speed clocked memories such as synchronous dynamicrandom access memory (SDRAM) devices.

Jitter, noise, and other factors sometimes interfere with the operationof DLL and PLL circuitry, so that the desired degree of synchronizationis not maintained. Thus, there is a need for improved apparatus,systems, and methods to improve periodic signal synchronization invarious electronic devices.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram of a DLL according to an embodiment of theinvention.

FIG. 2 is a diagram of a periodic feedback signal and a periodicreference signal associated with the DLL of FIG. 1 according to anembodiment of the invention.

FIG. 3 is a flow diagram of several methods associated with theoperation of the DLL in FIG. 1 according to an embodiment of theinvention.

FIG. 4 is a flow diagram of several methods associated with theoperation of the DLL in FIG. 1 according to an embodiment of theinvention.

FIG. 5 is a flow diagram of several methods associated with theoperation of the DLL in FIG. 1 according to an embodiment of theinvention.

FIG. 6 is a block diagram of a DLL including an electrical schematicdiagram of a circuit in the DLL according to an embodiment of theinvention.

FIG. 7 is a diagram of timing relationship signals associated with thecircuit of FIG. 6 according to an embodiment of the invention.

FIG. 8 is a diagram of voltages associated with the circuit of FIG. 6according to an embodiment of the invention.

FIG. 9 is a flow diagram of several methods associated with FIG. 6according to an embodiment of the invention.

FIG. 10 is a block diagram of a PLL according to an embodiment of theinvention.

FIG. 11 is a block diagram of a PLL according to an embodiment of theinvention.

FIG. 12 is a block diagram of a system according to an embodiment of theinvention.

FIG. 13 is a block diagram of a system according to an embodiment of theinvention.

DETAILED DESCRIPTION

FIG. 1 is a block diagram of a DLL 100 according to an embodiment of theinvention. An input buffer 110 is coupled to receive a periodic signalCK and an inverse periodic signal CKF. The signals CK and CKF arereceived from a source external to the DLL 100, and may be generated byan oscillator. The input buffer 110 generates a periodic referencesignal on a line 112 that is coupled to an interface control circuit114. The interface control circuit 114 is also coupled to receive afeedback signal on a line 116, and a generation of the feedback signalwill be discussed below.

The interface control circuit 114 couples the reference signal on a line118 to a coarse variable delay line 120, and couples the feedback signalon a line 119 and the reference signal on the line 118 to a phasedetection interface circuit 122. In some embodiments, the signals on thelines 118, 119 are clock signals. The interface control circuit 114 iscoupled to receive a control signal from a control logic circuit 124 toshut it off and save power when possible.

The coarse variable delay line 120 includes a series connection of aplurality of delay elements 125 that impart a delay to the referencesignal according to instructions from a shift register 126 coupled tothe coarse variable delay line 120. The shift register 126 containsbinary bits used to select an entry point for the reference signal intothe coarse variable delay line 120, and the entry point determines thenumber of delay elements that the reference signal is coupled through inthe coarse variable delay line 120. The coarse variable delay line 120generates a delayed reference signal on a pair of lines 127. A selectednumber of the delay elements in the coarse variable delay line 120 areincluded in a buffer 128, and these delay elements always impart delayto the reference signal.

The delayed reference signal on the lines 127 is coupled to a finevariable delay line 130 that imparts a further delay to the referencesignal based on instructions from the shift register 126. The finevariable delay line 130 generates an output signal on a line 132 that isfurther delayed from the reference signal according to the instructionsfrom the shift register 126. The binary bits in the shift register 126are used to select an entry point for the delayed reference signal inthe fine variable delay line 130. Each delay element in the coarsevariable delay line imparts a greater delay to the reference signal whencompared to the delay imparted by each delay element in the finevariable delay line 130.

An output buffer 134 couples the output signal to pins DQ and DQS (notshown). The output signal on a line 132 is also coupled through delaymodel circuit 136 that mimics a timing delay external to the DLL 100 inorder to generate a feedback signal on a line 138. The feedback signalon the line 138 is further coupled through a replica buffer circuit 140that is a replica of the input buffer 110 to impart a delay to thefeedback signal similar to the delay imparted by the input buffercircuit 110. The replica buffer circuit 140 generates the feedbacksignal on the line 116.

The phase detection interface circuit 122 detects a phase differencebetween the reference signal and the feedback signal on the lines 118and 119, respectively. Information about the phase difference in a phasedifference signal is coupled to a high speed multi measure logic circuit150 that includes a timing sequence controller circuit 152. A high speedclock generator circuit 154 generates a high speed clock signal that iscoupled to the phase detection interface circuit 122, the high speedmulti measure logic circuit 150, the timing sequence controller circuit152, and to an averaging initialization generator circuit 156. In someembodiments, the high speed multi measure logic circuit 150 and thetiming sequence controller circuit 152 are included in a digital signalprocessor (DSP). The high speed multi measure logic circuit 150 iscoupled to exchange information with the control logic circuit 124 overa line 157.

In some embodiments, the high speed clock generator circuit 154comprises a ring oscillator circuit. The high speed multi measure logiccircuit 150 and the timing sequence controller circuit 152 determine anaverage phase difference between the reference signal and the feedbacksignal on the lines 118, 119 based on multiple samples of the phasedifference from the phase detection interface circuit 122. Informationabout the average phase difference is coupled to the averaginginitialization generator circuit 156 which is coupled to the shiftregister 126 on line 158. The averaging initialization generator circuit156 is coupled to change binary bits in the shift register 126 tochange, in turn, the delay imparted by the coarse variable delay line120 and the fine variable delay line 130. Operation of the phasedetection interface circuit 122, the high speed multi measure logiccircuit 150, the timing sequence controller circuit 152, the high speedclock generator circuit 154, and the averaging initialization generatorcircuit 156 will be further described herein below.

FIG. 2 is a diagram of a periodic feedback signal 210 and a periodicreference signal 220 associated with the DLL 100 of FIG. 1 according toan embodiment of the invention. The feedback signal 210 corresponds tothe feedback signal on the lines 116,119 shown in FIG. 1, and thereference signal 220 corresponds to the reference signal on the lines112,118 shown in FIG. 1. The reference signal 220 and the feedbacksignal 210 include broken lines to indicate portions of the signal thatare repeated and not shown for purposes of brevity. A phase differenceΔT exists between the reference signal 220 and the feedback signal 210,and this phase difference ΔT can be detected for each period of thesignals 210, 220. For example, the phase difference ΔT is shown at Mdifferent intervals in FIG. 2, at ΔT₁, ΔT₂, and ΔT₃ through to ΔT_(M). Aphase difference between the feedback signal 210 and the referencesignal 220 is a delay interval between corresponding transitions (e.g.edges) of the signals 210, 220.

FIG. 3 is a flow diagram of several methods 300 associated with theoperation of the DLL 100 in FIG. 1 according to an embodiment of theinvention. The methods also express the operation of a PLL includingsimilar elements, and this PLL portion is described below.

The methods 300 start in block 310. In block 312, the DLL 100 isinitialized. In block 320, the reference signal on the line 112 beginsto be generated and clocked into the interface control circuit 114. Inblock 330, the feedback signal on the line 116 is detected, the highspeed clock generator circuit 154 begins to generate a high speed clocksignal, and a clocking generator period Tc of the feedback signal ismeasured.

In block 340, the phase detection interface circuit 122 is enabled todetect multiple phase differences ΔTi between the reference signal onthe line 118 and the feedback signal on the line 119 where i ranges from1 to an integer M. Also in block 340, a time-to-digital conversion ofthe phase differences ΔTi begins, and the phase differences ΔTi areconverted into digital data Di. In block 350, the phase detectioninterface circuit 122 is made ready to detect the next phase differenceΔTi, and the high speed multi-measure logic circuit 150 stores thecurrent value Di. In block 360, the methods 300 determine if M phasedifferences ΔT have been measured, and if not, the methods 300 return toblock 340 where another phase difference ΔTi is detected and convertedinto digital data Di.

If the methods 300 determine in block 360 that M samples ΔTi of thephase difference ΔT have been measured and converted into digital dataDi, then in block 370 the methods 300 calculate an average Davg of thesamples by summing the digital data Di, and dividing the sum by theinteger M. Also in block 370, the averaging initialization generatorcircuit 156 generates an average pulse width based on the average Davgof the phase difference. In block 380, the averaging initializationgenerator circuit 156 updates the shift register 126 with new binarybits to adjust the phase locking of the variable delay lines 120,130. Inparticular, an enable token is sent through the variable delay lines120,130 starting from a beginning edge of the average pulse width andstopping on a disable edge of the average pulse width. Shift controllogic (not shown) is then used to latch in or register binary bits inthe shift register 126 based on the enable token. In block 382, a normaloperation mode begins, and in block 384, the methods 300 end.

FIG. 4 is a flow diagram of several methods 400 associated with theoperation of the DLL 100 in FIG. 1 according to an embodiment of theinvention. FIG. 5 is a flow diagram of several methods 500 associatedwith the operation of the DLL 100 in FIG. 1 according to an embodimentof the invention, and will be described along with the methods 400. Thatis, the methods 400 and 500 show the operation of the DLL 100 in someembodiments distinguished from the methods 300 shown in FIG. 3.

The methods 400 start in block 410. In block 420, an integer M isselected to determine a number of samples of a phase difference thatwill be measured in the methods 400. In block 430, a feedback loopincluding the delay model circuit 136 and the replica buffer circuit 140is reset, and a reference clock signal on the line 112 is coupled to theinterface control circuit 114. In block 440, a feedback clock signal onthe line 116 is detected, and the high speed clock generator circuit 154is enabled to generate a multi-phase clock signal having N phases, whereN is an integer, and each phase has a period approximately equal to aperiod of the reference clock signal on the line 118. The phase of thereference clock signal on line 118 is Tclk, and a phase difference Δtbetween each of the N multi phase clock signals is Tclk divided by N.

In block 450, the phase detection interface circuit 122 is enabled todetect a phase difference between the reference clock signal on the line118 in the feedback clock signal on the line 119. Also in block 450, thehigh speed multi measure logic circuit 150 and the timing sequencecontroller circuit 152 are enabled to carry out a time-to-digitalconversion of the phase difference based on the multi-phase clock signalgenerated by the high speed clock generator circuit 154.

The coarse value and the fine value of the phase difference aredetermined in the following manner. The measurement of a phasedifference between an edge of the reference clock signal on the line 118and an edge of the feedback clock signal on the line 119 is carried outby a comparison with the N multi-phase clock signals. Between edges ofthe signals on the lines 118, 119, a phase difference detector for eachmulti-phase clock signal detects when an edge of that multi-phase clocksignal occurs, and a counter is incremented by one. There are N phasedifference detectors and N counters that count up during the phasedifference. The N counters each begin at 0. When the edges defining thephase difference have passed, the data in the counters is used to findthe length of the phase difference. A Coarse value is the value in thecounter. A Fine value is the number of counters having the highest ormaximum Coarse value. The duration (e.g. Time) of the phase differenceis calculated by formula (I) where * indicates multiplication:

Time=(Coarse−1)*Tclk+Fine*Δt  (1)

In block 460, the maximum Coarse value and a Fine value of distanceequivalency of a sample are determined. The distance equivalency is thenumber of counters having the same value. Also in block 460, the phasedetection interface circuit 122 is disabled. In block 470, the methods400 determine if M different samples of the phase difference have beenmeasured, and if not, the methods return to the block 450 where thephase detection interface circuit 122 is enabled and another sample ofthe phase difference is measured. Each time the maximum Coarse value andthe Fine value of distance equivalency are determined in block 460,those values are stored in block 480. If, in block 470, the methods 400determine that M samples of the phase difference have been measured, thehigh speed multi measure logic circuit 150 performs DSP averaging inblock 480 according to the methods 500 shown in FIG. 5.

In block 510 of FIG. 5, the methods 500 begin. In block 520, there is aninitialization start and two average variables Qc_avg and Qf_avg are setto zero. In block 530, the Coarse values Qcoarse are summed to obtain avariable Qc_sum, and the Fine values Qfine are summed to obtain avariable Qf_sum. In block 540, interaction terms and temporary variablesare calculated according to the following equations. In these equations,the modulo function (mod) is used to find the remainder of division ofone number by another. Z mod M is the remainder of a division of Z by M.

Qc2f=N*mod(Qc_sum,M)  (2)

Qf_tmp=[{Qf_sum+Qc2f+0 . . . 010}/M]  (3)

Qf2c=[Qf_tmp/N]  (4)

In these equations, C=N*f. In block 550, average values are thencalculated according to the following formulas:

Qf_avg=mod(Qf_tmp,N)  (5)

Qc_avg=[Qc_sum/M]+Qf2c  (6)

In block 560, the average values are updated to the averaginginitialization generator circuit 156, and in block 570, the averagingDSP is complete. The averaging in the methods 400 and 500 is carried outby a DSP. In block 580, the methods 500 end.

Returning to the methods 400 shown in FIG. 4, in block 490 multi-phaseclocking DSP decoder high speed average initialization pulse generationis carried out by the averaging initialization generator circuit 156.The averaging initialization generator circuit 156 generates an averagepulse width based on the average values Qf_avg and Qc_avg of the phasedifference using the same multi-phase clock that was used to sample thephase differences. The use of the same multi-phase clock should improvethe fidelity of the digital method.

In block 492, the averaging initialization generator circuit 156 updatesthe shift register 126 with new binary bits to adjust the phase lockingof the variable delay lines 120,130 of the DLL 100. In particular, anenable token is sent through the variable delay lines 120,130 startingfrom a beginning edge of the average initialization pulse and stoppingon a disable edge of the average initialization pulse. Shift controllogic (not shown) is then used to latch in or register binary bits inthe shift register 126 based on the enable token. In block 494, a normaloperation mode of the DLL 100 begins, and in block 496 the methods 400end.

FIG. 6 is a block diagram of a DLL 600 including an electrical schematicdiagram of a circuit in the DLL 600 according to an embodiment of theinvention. The DLL 600 includes many elements similar to the DLL 100shown in FIG. 1, and such similar elements have been given the samereference numerals and will not be described further herein for purposesof brevity. FIG. 7 is a diagram of timing relationship signalsassociated with the circuit of FIG. 6 according to an embodiment of theinvention. FIG. 8 is a diagram of voltages associated with the circuitof FIG. 6 according to an embodiment of the invention.

A phase detection interface circuit 650 is coupled to receive thereference signal on the line 118 and the feedback signal on the line119, and to generate a voltage Va on a line 652, shown in FIG. 7, thatis high during a phase difference between the reference signal on theline 118 and the feedback signal on the line 119. The voltage Va may becalled a phase difference signal. An operational amplifier 660 operatesas a full swing comparator between an average capacitance Cavg 662coupled to an inverting input of the operational amplifier 660, and alarger capacitance Csum 664 coupled to a non-inverting input of theoperational amplifier 660. The larger capacitance Csum is approximatelyequal to an integer M multiple of the smaller capacitance Cavg. Theoperational amplifier 660, the capacitance Cavg 662, and the capacitanceCsum 664 are included in an analog circuit used to determine an averageof a phase difference between the reference signal on the line 118 andthe feedback signal on the line 119 based on M samples of the phasedifference.

An output of the operational amplifier 660 is coupled to an input of aninverter 670, and an output of the inverter 670 is coupled to anaveraging pulse Vb circuit 672 and to a disable switch 674 as will bedescribed below. A switch K₁ 680 is coupled between a current source682, a first node a coupled to the larger capacitance Csum and a secondnode b coupled to the smaller capacitance Cavg 662. The switch K₁ 680toggles back and forth between the node a and the node b as will bedescribed below. The averaging pulse Vb circuit 672 is coupled toexchange information with the control logic circuit 124 over a line 683.The control logic circuit 124 is also coupled to send information to theshift register 126 over a line 684, and is coupled to send informationto the phase detection interface circuit 650 over a line 686.

FIG. 7 shows the voltage V_(a) on the line 652 which is high duringintervals of a phase difference between the reference signal on the line118 and the feedback signal on the line 119. A voltage V_(b) indicatesthe voltage on the node b that will be described below. A pulse is shownthat controls a switch K₀ 668 that couples the inverting andnon-inverting inputs of the operational amplifier 660 when the pulse ishigh. Also shown in FIG. 7 is an output signal V_(out) on the output ofthe operational amplifier 660.

As shown in FIG. 8, initially, the switch K₀ 668 couples the invertingand non-inverting inputs of the operational amplifier 660 such that asubstantially negligible voltage difference occurs between them before atime T_(a). When the voltage V_(a) is high, the switch K₁ 680 is coupledbetween the current source 682, the node a, and the larger capacitanceCsum to charge the capacitance Csum. As shown in FIG. 8 between timeT_(a) and time T_(b), a potential on the larger capacitance Csum riseswhenever V_(a) is high during a phase difference between the referencesignal on the line 118 and the feedback signal on the line 119. Thelarger capacitance Csum 664 is charged M times before it reaches amaximum voltage and at the end of M samples of the phase difference.

In FIG. 8, the voltage V_(a) goes low after the capacitance Csum 664 ischarged M times, and, after a suitable delay, the voltage V_(b) iscontrolled to go high at a time T_(b) to couple the switch K₁ 680between the current source 682 and the node b to charge the smallercapacitance Cavg 662. The capacitance Cavg 662 has a voltage that risesfaster than the voltage on the larger capacitance Csum 664 as is shownin FIG. 8. When a potential on the smaller capacitance Cavg exceeds thepotential on the larger capacitance Csum, the operational amplifier 660generates a low output signal V_(out) coupled to the input of theinverter 670. The inverter 670 then generates a high signal to cause thedisable switch 674 to disable the switch K₁ 680 and to indicate to theaveraging pulse V_(b) circuit 672 that the capacitance Cavg is charged.The interval between the rise of the voltage V_(b) and the fall of theoutput signal V_(out) is approximately equal to the average phasedifference between the reference signal on the line 118 and the feedbacksignal on the line 119, and is shown as ΔT_(avg) in FIG. 8. The voltageV_(b) is brought low following the fall of the output signal V_(out).

The averaging pulse V_(b) circuit 672 generates an averageinitialization pulse having a duration approximately equal to theinterval between the rise of the voltage V_(b) and the fall of theoutput signal V_(out). The average initialization pulse is coupled to anaveraging initialization generator circuit 690. The averaginginitialization generator circuit 690 sends an enable token through thevariable delay lines 120,130 starting from a beginning edge of theaverage initialization pulse and stopping on a disable edge of theaverage initialization pulse. Shift control logic (not shown) is thenused to latch in or register binary bits in the shift register 126 basedon the enable token to latch a new delay into the DLL 600.

FIG. 9 is a flow diagram of several methods 900 associated with FIG. 6according to an embodiment of the invention. The methods 900 generallyexpress the operation of the DLL 100. The methods also express theoperation of a PLL including similar elements and this PLL portion isdescribed herein below.

The methods 900 start in block 910. In block 920, the DLL 600 isinitialized. In block 930, the reference signal on the line 112 beginsto be generated and clocked into the interface control circuit 114. Inblock 940, the feedback signal on the line 116 is detected. In block950, the phase detection interface circuit 650 is enabled to detectmultiple phase differences ΔTi between the reference signal on the line118 and the feedback signal on the line 119 where i ranges from 1 to aninteger M. The phase detection interface circuit 650 is enabled togenerate the voltage V_(a) such that the switch K₁ 680 is coupled tocharge the larger capacitance Csum. In block 960, the phase detectioninterface circuit 650 is made ready to detect the next phase differenceΔTi. In block 970, the methods 900 determine if M phase differences ΔThave been measured, and if not, the methods 900 return to block 950where another phase difference ΔTi is detected.

If the methods 900 determine in block 970 that M samples ΔTi of thephase difference ΔT have been measured and the capacitance Csum 664 ischarged M times, then in block 980 the switch K₁ 680 is coupled to thenode b to charge the smaller capacitance Cavg. Once the capacitance Cavgis charged, the disable switch 674 is caused to disable the switch K₁680. In block 990, the averaging initialization generator circuit 690updates the shift register 126 with new binary bits to adjust the phaselocking of the variable delay lines 120,130. In block 992, a normaloperation mode begins, and in block 994, the methods 900 end.

FIG. 10 is a block diagram of a PLL 1000 according to an embodiment ofthe invention. The PLL 1000 includes a phase detector 1010, an averagingloop filter 1020, a voltage controlled oscillator (VCO) 1030, and adivide-by-N counter 1040. The PLL 1000 also includes digital circuitssimilar to the circuits shown in FIG. 1, including the high speedmulti-measure logic circuit 150, the timing sequence controller circuit152, and the high speed clock generator circuit 154. The circuits 150,152, and 154 are given the same reference numerals in the PLL 1000, andoperate in a manner similar to the operation described with reference toFIG. 1; this operation will not be further described for purposes ofbrevity.

The PLL 1000 aligns a rising edge of a reference signal on a line 1052to a feedback signal on a line 1054 coupled to the phase detector 1010.The VCO 1030 oscillates to generate an output signal on a line 1056 at afrequency that determines the phase and frequency of the feedback signalon the line 1054. The phase detector 1010 detects a phase differencebetween the reference signal on the line 1052 and a feedback signal onthe line 1054. Information about the phase difference is coupled in aphase difference signal to the high speed multi measure logic circuit150 over lines 1060.

The high speed multi-measure logic circuit 150 generates informationbased on an average of multiple samples of the phase difference betweenthe signals on the lines 1052 and 1054, and exchanges this informationwith the averaging loop filter 1020 over lines 1070. Based on thisinformation, the averaging loop filter 1020 determines whether the VCO1030 needs to operate at a higher or lower frequency, and generates acontrol voltage on a line 1080 that is coupled to bias the voltagecontrolled oscillator 1030. The VCO 1030 oscillates to generate theoutput signal on the line 1056, and stabilizes once the reference signalon the line 1052 and the feedback signal on the line 1054 have the samephase and frequency. When the reference signal and the feedback signalare aligned, the PLL 1000 is considered locked. The divide-by-N counter1040 increases the frequency of the VCO 1030 above the frequency of thereference signal on the line 1052.

The methods 300 shown in FIG. 3 are altered slightly according to thestructure of the PLL 1000 shown in FIG. 10. In block 312, the PLL 1000is initialized (e.g. reset). In block 320, the reference signal on theline 1052 begins to be generated and clocked into the phase detector1010. In block 330, the feedback signal on the line 1154 is detected. Inblock 380, the input of the VCO 1030 is adjusted by a linear transferfunction of k*ΔDavg where k is a fine-tune factor in the PLL 1000 systemconfiguration.

FIG. 11 is a block diagram of a PLL 1100 according to an embodiment ofthe invention. The PLL 1100 includes a phase detector 1110, a chargepump 1116, a loop filter 1120, a VCO 1130, and a divide-by-N counter1140. The PLL 1100 also includes analog circuits similar to the circuitsshown in FIG. 6, including the operational amplifier 660, the averagecapacitance Cavg 662, and the larger capacitance Csum 664. The circuitelements shown in FIG. 6 are given the same reference numerals in thePLL 1100, and operate in a manner similar to the operation describedwith reference to FIG. 6; this operation will not be further describedfor purposes of brevity.

The PLL 1100 aligns a rising edge of a reference signal on a line 1152to a feedback signal on a line 1154 coupled to the phase detector 1110.The VCO 1130 oscillates to generate an output signal on a line 1156 at afrequency that determines the phase and frequency of the feedback signalon the line 1154. The phase detector 1110 detects a phase differencebetween the reference signal on the line 1152 and a feedback signal onthe line 1154. The phase detector 1110 generates the voltage V_(a) on aline 1160 based on information about the phase difference. The voltageV_(a) may be called a phase difference signal. The averaging pulse V_(b)circuit 672 generates an up signal on a line 1162 or a down signal on aline 1164 coupled to the charge pump 1116 based on an average ofmultiple samples of the phase difference between the signals on thelines 1152 and 1154. If the charge pump 1116 receives an up signal,current is driven into the loop filter 1120 on a line 1170. Conversely,if the charge pump 1116 receives a down signal, current is drawn fromthe loop filter 1120 on the line 1170.

The loop filter 1120 converts these signals to a control voltage on aline 1180 that is used to bias the VCO 1130. Based on the controlvoltage, the VCO 1130 oscillates at a higher or lower frequency togenerate the output signal on the line 1156 which affects the phase andfrequency of the feedback signal on the line 1154. The VCO 1130stabilizes once the reference signal on the line 1152 and the feedbacksignal on the line 1154 have the same phase and frequency. When thereference signal and the feedback signal are aligned, the PLL 1100 isconsidered locked. The divide-by-N counter 1140 increases the frequencyof the VCO 1130 above the frequency of the reference signal on the line1152.

The methods 900 shown in FIG. 9 are altered slightly according to thestructure of the PLL 1100 shown in FIG. 11. In block 920, the PLL 1100is initialized (e.g. reset). In block 930, the reference signal on theline 1152 begins to be generated and clocked into the phase detector1110. In block 940, the feedback signal on the line 1154 is detected. Inblock 990, the input of the VCO 1130 is adjusted by a linear transferfunction of k*ΔTavg, where k is a fine-tune factor in the PLL 1100system configuration.

Embodiments of the invention described herein determine an average phasedifference between a periodic output signal and a periodic input signalfrom an average of a plurality of samples of a phase difference betweenthe output signal and the input signal. A phase of the output signal isthen adjusted based on the average phase difference.

In some embodiments of the invention, the phase of the output signal maybe adjusted based on other mathematical treatment of the samples of thephase difference. In some embodiments of the invention, a calculatedphase difference may be calculated by alternately adding and subtractingsucceeding samples of the phase difference. For example, the calculatedphase difference may be equal to a phase difference of sample A−a phasedifference of sample B+a phase difference of sample C−a phase differenceof sample D+a phase difference of sample E. In some embodiments of theinvention, the signs in the above equation are reversed, or more samplesare included in the calculation.

In some embodiments of the invention, the calculated phase differencemay be calculated by adding or subtracting differently weighted samplesof the phase difference. For example, the calculated phase differencemay be equal to k1*phase difference of sample A+/−k2*phase difference ofsample B+/−k3*phase difference of sample C where k1, k2, and k3 aredifferent weighting factors. In some embodiments of the invention, thesigns in the above equation are reversed, or more samples are includedin the calculation. In some embodiments of the invention, the calculatedphase difference may be calculated from a least variance approximationof the samples of the phase difference. In some embodiments of theinvention, the calculated phase difference may be calculated from aleast squares estimate of the samples of the phase difference. In someembodiments of the invention, the calculated phase difference may becalculated from an Nth root of a product of the samples of the phasedifference.

Embodiments of the invention described herein may be included inelectronic circuitry used in high-speed computers, communication andsignal processing circuitry, single or multi-processor modules, singleor multiple embedded processors, multi-core processors, data switches,and application-specific modules including multilayer, multi-chipmodules. Such apparatus and systems may further be included assub-components within a variety of electronic systems, such astelevisions, cellular telephones, personal computers (e.g., laptopcomputers, desktop computers, handheld computers, tablet computers,etc.), workstations, radios, video players, audio players (e.g., MP3(Motion Picture Experts Group, Audio Layer 3) players), vehicles,medical devices (e.g., heart monitor, blood pressure monitor, etc.), settop boxes, and others. Thus, many more embodiments may be realized, someof which are described below.

FIG. 12 is a block diagram of a system 1200 according to an embodimentof the invention. The system 1200 includes a processor 1202 and a memorydevice 1203 that includes an input/output (I/O) control block 1204, acontrol logic block 1206, a pipelined latches and registers block 1210,and a memory array 1211. In some embodiments, the memory array 1211comprises a NAND flash memory array, a NOR flash memory array, or aSDRAM array.

The memory device 1203 includes a DLL or a PLL 1212 to generate aperiodic signal such as a clock signal according to embodiments of theinvention described herein.

In some embodiments, the processor 1202 and the memory device 1203 maybe included on a single integrated circuit. In an embodiment of theinvention, the memory device 1203 may be included as removable storagesuch as flash cards and USB Flash drives, and may be included asembedded storage for cell phones, digital cameras, wireless/handhelddevices, and MP3 players.

The system 1200 includes column decoders 1214, row decoders 1216, and adata registers block 1218. The column decoders 1214 are coupled to thememory array 1211 and provide column selection signals. The row decoders1216 are coupled to the memory array 1211 and provide row selectionsignals. The data registers block 1218 is coupled to the memory array1211. The data registers block 1218, in an embodiment of the invention,includes one or more data registers for transferring data to and fromthe memory array 1211. In some embodiments, the data registers block1218 may include cache registers 1219.

In the system 1200, the processor 1202 may be coupled to the I/O controlblock 1204 thorough an interconnect 1220 and the control logic block1206 through an interconnect 1222. The I/O control block 1204 may becoupled to the control logic block 1206 through an interconnect 1223.The pipelined latches and registers block 1210 may be coupled to the I/Ocontrol block 1204 through an interconnect 1224 and to the control logicblock 1206 through an interconnect 1226. The pipelined latches andregisters block 1210 may be connected through an interconnect 1228 tothe column decoders 1214, the row decoders 1216, and to the dataregisters 1218. In some embodiments, the data registers 1218, includingthe cache registers 1219 if present, may be coupled to the I/O controlblock 1204 through and interconnect 1230.

The interconnects 1220, 1222, 1223, 1224, 1226, 1228, and 1230 are notlimited to any particular type of interconnects. In some embodiments,one or more of these interconnects may include a plurality of individualconductors operating in parallel to transfer data. Serial connectionsmay also be made. In some embodiments, one or more of theseinterconnects may include a wireless interconnect. The interconnects1220, 1222, 1223, 1224, 1226, 1228, and 1230 are not limited to beingthe same type of interconnects, and the system 1200 may include avariety of interconnects 1220, 1222, 1223, 1224, 1226, 1228, and 1230,perhaps used in combination.

In operation, the processor 1202 provides a plurality of control signalsthrough the interconnect 1222 to the control logic block 1206 to controloperations performed on the memory array 1211. Operations the processor1202 performs in controlling the memory array 1211 include, but are notlimited to, reading data from the memory array 1211, writing data to thememory array 1211, and erasing one or more portions of the memory array1211. In order to perform an operation, the processor 1202 provides tothe I/O control block 1204 a series of command signals, address signals,and data signals through the interconnect 1220, and the command signals,the address signals, and the data signals are all used in performing theoperation.

In some embodiments, the command signals, the address signals, and thedata signals may be sent as a series of serial bytes from the processor1202 though the interconnect 1220 to the I/O control block 1204, and arelatched into latches included in the pipelined latches and registersblock 1210. In some embodiments, the command signals may be decoded andprovided to the control logic block 1206 for the control logic block1206 to generate internal signals for controlling the operation beingperformed on the memory array 1211. Further, the address signals may belatched into the pipelined latches and registers block 1210, andprovided to the column decoder 1214 and to the row decoder 1216 tocontrol a portion of the memory array 1211 on which an operation isbeing performed. Data signals are latched into registers of thepipelined latches and registers block 1210 and are provided to dataregisters 1218, for example, during write operations to the memory array1211.

During operations on the memory array 1211, the processor 1202 may alsoprovide one or more control signals through the interconnect 1222 to thecontrol logic block 1206. These control signals are provided andcoordinated with the signals provided by the processor 1202 to the I/Ocontrol block 1204 to control the operations performed on the memoryarray 1211. In addition, the control logic block 1206 may provide one ormore internally generated control signals to the I/O control block 1204and to the pipelined latches and registers block 1210 to control theoperations performed on memory array 1211.

FIG. 13 is a block diagram of a system 1300 according to an embodimentof the invention. The system 1300 may include a processor 1310, an imagesensor device 1320, a memory device 1325, a memory controller 1330, agraphics controller 1340, a circuit module 1345, an I/O controller 1350,a display 1352, a keyboard 1354, a pointing device 1356, a peripheraldevice 1358, and a bus 1360 to transfer information among the componentsof system 1300. The system 1300 may also include a circuit board 1302 onwhich some components of the system 1300 may be located. In someembodiments, the number of components of system 1300 may vary. Forexample, in some embodiments, the system 1300 may omit one or more ofthe display 1352, the image sensor device 1320, the memory device 1325,and the circuit module 1345.

The memory device 1325 includes a DLL or a PLL 1370 to generate aperiodic signal such as a clock signal according to embodiments of theinvention described herein. One or more of the processor 1310, the imagesensor device 1320, the memory controller 1330, the graphics controller1340, the circuit module 1345, the I/O controller 1350, the display1352, the keyboard 1354, the pointing device 1356, and the peripheraldevice 1358 may also include a DLL or a PLL to generate a periodicsignal such as a clock signal according to embodiments of the inventiondescribed herein.

The processor 1310 may include a general-purpose processor or anapplication specific integrated circuit (ASIC). The processor 1310 maycomprise a single core processor or a multiple-core processor. Theprocessor 1310 may execute one or more programming commands to processinformation to provide processed information. The information mayinclude digital output information provided by other components of thesystem 1300, such as the image sensor device 1320 or the memory device1325.

The image sensor device 1320 may include a complementarymetal-oxide-semiconductor (CMOS) image sensor having a CMOS pixel arrayor a charge-coupled device (CCD) image sensor having a CCD pixel array.

The memory device 1325 of FIG. 13 may include a volatile memory device,a non-volatile memory device, or a combination of both. For example, thememory device 1325 may comprise a DRAM device, a static random accessmemory (SRAM) device, a flash memory device, or a combination of thesememory devices.

The display 1352 may include an analog display or a digital display. Thedisplay 1352 may receive information from other components. For example,the display 1352 may receive information that is processed by one ormore of the image sensor device 1320, the memory device 1325, thegraphics controller 1340, and the processor 1310 to display informationsuch as text or images.

The circuit module 1345 may include a circuit module of a vehicle. Thecircuit module 1345 may receive information from other components toactivate one or more subsystem of the vehicle. For example, the circuitmodule 1345 may receive information that is processed by one or more ofthe image sensor device 1320, the memory device 1325, and the processor1310 to activate one or more of an air bag system of a vehicle, avehicle security alarm, and obstacle alert system.

The individual activities of methods 300, 400, 500, and 900 may not haveto be performed in the order shown or in any particular order. Someactivities may be repeated, and others may occur only once. Embodimentsof the invention may have more or fewer activities than those shown inFIGS. 3, 4, 5, and 9.

Any of the circuits or systems described herein may be referred to as amodule. A module may be a circuit or firmware according to embodimentsof the invention.

The above description and the drawings illustrate some embodiments ofthe invention to enable those skilled in the art to practice theembodiments of the invention. Other embodiments may incorporatestructural, logical, electrical, process, and other changes. In thedrawings, like features or like numerals describe substantially similarfeatures throughout the several views. Examples merely typify possiblevariations. Portions and features of some embodiments may be includedin, or substituted for, those of others. Many other embodiments will beapparent to those skilled in the art upon reading and understanding theabove description. Therefore, the scope of an embodiment of theinvention of the invention is determined by the appended claims, alongwith the full range of equivalents to which such claims are entitled.

The Abstract is provided to comply with 37 C.F.R. §1.72(b) requiring anabstract that will allow the reader to quickly ascertain the nature ofthe technical disclosure. The Abstract is submitted with theunderstanding that it will not be used to interpret or limit the scopeor meaning of the claims.

1. A method, comprising: generating a periodic output signal from aperiodic input signal; obtaining a plurality of samples of a phasedifference between the output signal and the input signal; determiningan average phase difference between the output signal and the inputsignal from the plurality of samples; and adjusting a phase of theoutput signal based on the average phase difference.
 2. The method ofclaim 1, wherein adjusting a phase of the output signal includesadjusting the phase of the output signal to establish the phasedifference between the output signal and the input signal ofapproximately zero.
 3. The method of claim 1, wherein adjusting a phaseof the output signal includes delaying the input signal based on theaverage phase difference to generate the output signal.
 4. The method ofclaim 1, wherein generating a periodic output signal includes generatinga periodic output clock signal.
 5. The method of claim 1, whereinadjusting a phase of the output signal further includes adjusting thephase of the output signal to maintain a predetermined phaserelationship between the output signal and the input signal.
 6. A methodcomprising: generating a periodic output signal from a periodic inputsignal; determining an average phase difference between the outputsignal and the input signal over a plurality of periods of the inputsignal; and delaying the input signal based on the average phasedifference to generate the output signal.
 7. The method of claim 6,wherein determining an average phase difference includes: charging afirst capacitance during each interval of a phase difference between theoutput signal and the input signal over the plurality of periods;charging a second capacitance over a charge interval until a potentialacross the second capacitance exceeds a potential across the firstcapacitance, the first capacitance being an integer multiple of thesecond capacitance; and determining the average phase difference to bethe charge interval over which the second capacitance was charged. 8.The method of claim 6, wherein delaying the input signal includes:shifting one or more bits in a shift register in response to the averagephase difference; and coupling the input signal to a variable delay lineat an entry point selected by the shift register to delay the inputsignal and to generate the output signal.
 9. A method comprising:generating a periodic output signal from a periodic input signal;obtaining a plurality of samples of a phase difference between theoutput signal and the input signal; and adjusting a phase of the outputsignal based on a combination of the samples of the phase difference.10. The method of claim 9, further comprising: determining an averagephase difference between the output signal and the input signal from anaverage of the plurality of samples, wherein adjusting a phase of theoutput signal includes adjusting the phase of the output signal based onthe average phase difference.
 11. The method of claim 9, whereinadjusting a phase of the output signal includes delaying the inputsignal based on the combination of the samples of the phase differenceto generate the output signal.
 12. The method of claim 9, whereinadjusting a phase of the output signal includes increasing or decreasinga delay between a transition of the output signal and a correspondingtransition of the input signal to establish a phase relationship betweenthe output signal and the input signal.
 13. The method of claim 9,wherein adjusting a phase of the output signal includes adjusting aphase of the output signal such that the output signal and the inputsignal are in phase.
 14. An apparatus comprising: a feedback loop togenerate a periodic output signal from a periodic input signal; a firstcapacitance to be charged by a current source during an integer numberof intervals of a phase difference between the input signal and theoutput signal; a second capacitance approximately equal to the firstcapacitance divided by the integer, the second capacitance to be chargedby the current source until a potential across the second capacitanceexceeds a potential across the first capacitance, an average phasedifference between the input signal and the output signal beingapproximately equal to a time during which the second capacitance wascharged by the current source; and a module to adjust a phase of theoutput signal based on the average phase difference to establish a phaserelationship between the output signal and the input signal.
 15. Theapparatus of claim 14, further comprising: a comparator having aninverting input coupled to the second capacitance, a non-inverting inputcoupled to the first capacitance, and an output to generate a disablesignal when the potential across the second capacitance exceeds thepotential across the first capacitance; a charging switch to couple thecurrent source to the first capacitance during the intervals of thephase difference between the input signal and the output signal and tocouple the current source to the second capacitance until the potentialacross the second capacitance exceeds the potential across the firstcapacitance; a disable switch coupled to the output of the comparatorthrough an inverter to disable the charging switch in response to thedisable signal; and a digital circuit coupled to the output of thecomparator to generate a pulse in response to the disable signal, awidth of the pulse being based on the time during which the secondcapacitance was charged by the current source.
 16. The apparatus ofclaim 14, wherein the feedback loop comprises a phase lock loopincluding an oscillator.
 17. The apparatus of claim 15, wherein thefeedback loop comprises a delay lock loop including a variable delayline and a shift register coupled to the digital circuit, the variabledelay line being coupled to the input signal at a location selected bythe shift register to delay the input signal to generate the outputsignal.
 18. A system comprising: a processor coupled to a bus; anelectronic device, the electronic device including a feedback loop togenerate a periodic internal signal from a periodic external signalreceived from the bus, the feedback loop including a circuit to measurean average phase difference between the internal signal and the externalsignal based on a plurality of samples of the phase difference, and toadjust a phase of the internal signal based on the average phasedifference.
 19. The system of claim 18, wherein the electronic device isselected from the group consisting of one or more of a processor, animage sensor device, a display, a memory device, a memory controller, agraphics controller, a circuit module, an input and output controller, akeyboard, a pointing device, or a peripheral device.
 20. The system ofclaim 18, wherein: the external signal comprises an external clocksignal; and the internal signal comprises an internal clock signal. 21.The system of claim 18, wherein the feedback loop comprises a delay lockloop.
 22. The system of claim 18, wherein the feedback loop comprises aphase lock loop.
 23. The system of claim 18, wherein the feedback loopincludes: a clock generator to generate a selected clock signal; a phasedetector having a first input coupled to receive the external signal, asecond input coupled to receive the internal signal, and an output, thephase detector to generate a phase difference signal on the outputindicating a phase relationship between the external signal and theinternal signal; a digital circuit coupled to the output of the phasedetector to receive the phase difference signal and coupled to the clockgenerator to receive the selected clock signal, the digital circuit tocompare the phase difference signal with the selected clock signal togenerate a plurality of phase difference measurements over a pluralityof periods of the input signal, and to calculate an average phasedifference from an average of the phase difference measurements; and anoutput circuit coupled to the digital circuit to adjust a phase of theinternal signal based on the average phase difference.
 24. The system ofclaim 23, wherein the clock generator is coupled to generate amulti-phase clock signal having a plurality of phases, each phase of themulti-phase clock signal having a period approximately equal to a periodof the external signal.
 25. The system of claim 18, wherein the feedbackloop includes: a first capacitance to be charged by a current sourceduring an integer number of intervals of a phase difference between theexternal signal and the internal signal; a second capacitanceapproximately equal to the first capacitance divided by the integer, thesecond capacitance to be charged by the current source until a potentialacross the second capacitance exceeds a potential across the firstcapacitance, an average phase difference between the external signal andthe internal signal being approximately equal to a time during which thesecond capacitance was charged by the current source; and a circuit toadjust a phase of the internal signal based on the average phasedifference to establish a phase relationship between the internal signaland the external signal.